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Everything posted by AN920

  1. Do you mean, (serie-shunt,shunt-shunt,series-series and shunt-series)? http://www.ece.mtu.edu/faculty/goel/EE-4232/Feedback.pdf http://www.scribd.com/doc/92675/EE3102-Experiment-1
  2. Here are some pictures showing the 100 kHz noise pulses into the VCO tuning line. In the first picture it is not easy to spot the problem. The scale is 1mV/div The next picture shows the pulses after I averaged it 64 times. The 100 kHz pulses can be clearly seen. These pulses are very narrow. You won't even see them on a 20 MHz BW scope.
  3. You need to provide more information 1) Which IGBT's are you using? 2) What size motor are you driving? 3) What type of load. Pump, fan etc. Under which conditions do they fail?
  4. Hi Sarma, No, that won't help as I will show below. If you look at the first plot, you will see that with the 47nF connected between the base-collector of the MPSA13 the gain at 100 KHz is about 6.5dB. This is the problem I suspected. In the second plot, by moving the 47nF to the base-emitter of the MPSA13 you make the problem much worse. Now you will have 14.5dB at 100 kHz. No good. The last plot shows putting the capacitor at the 2nd base of the pair of 2N3904's. I increased the value from my previous suggestion after doing a simulation to 22nF. Now the gain at 100 KHz is zero. With my original value of 10nF there was still about 1dB gain at 100 kHz.
  5. A good friend of mine in Australia bought one of these kits. He was having some problems getting it to work properly. He posted it to me to look at. Here are my findings and some modifications I did. I will address it point by point: 1) Move this cap closer to pin 12. The PLL Vcc is very sensitive to any noise. (see pcb layout below) 2) Run the VCO tuning supply from a higher voltage than 5V. It must be regulated. This can be another small 78L09 or suitable zener. This will help locking at bottom and top ends of the band without readjusting L1. 3) Reduce the 5.1k to about 2.7k (It will allow faster tuning). The 22k On the base of Q1 can also be made slightly smaller with not much effect on performance. 4) Try a variable capacitor there to alter the phase of the pilot for best stereo separation. You can replace it with a fixed value after measuring on a LCR meter. 5) Try to get the ratio of the (22pF, 33pF and VC1 to 10pF) as high as possible. This means the total capacitance by the 3 elements in the circle should be as large as possible while the series cap (10pF in this case) should be as small as possible. L1 should be as small as possible. This may take some experimentation to get values that will provide lock at the top and bottom end. If you operate on a single frequency this is not a problem. This will give lowest carrier noise sensitivity for this clapp oscillator. 6,7) I found some slight loop instability (hunting) when tuned to the high end of the band. This sounded like a faint whistle on the audio. I think it is caused by some of the 100 kHz comp reference still getting into the VCO tuning line from the output of the charge pump. I solved this by using 2 separate transistors for Q1. This gave me access to the base of the second transistor. I removed the 47nF and placed a smaller 10nF (make this 22nF, see next post) as shown. This cured the instability and noise problem. When you set up your transmitter at the low end and your voltage at TP1 reads lower than about 0.7V or is stuck at 0.6V then the VCO can
  6. Someone suggested the faster LT1353 to extend the operation to over 100 KHz. I will look into this. The PIC clock will have to be much faster for the counter.
  7. The current probe or current transformer method was suggested by Dr. Middlebrook long ago. Every injection point has its pros and cons. I think where a loop is very marginally stable various methods may make a small difference, but where it shows obvious instability it won't make much of a difference. By using the better driver we can get it unconditionally stable, so why not? The price is only $0.18 Someone who tests the supply for load switching may see no oscillation, because of high ESR output cap. He will see a voltage dip and overshoot when the load is removed. The real test will be with a high grade (switchmode type) electrolytic with low ESR.
  8. When I started to fly RC planes long ago (I don't anymore), I started with a STIK-40 with a tuned pipe. Radios with 4-6 channels were expensive. There were many beginners who had 2-chan radios. When the local club found out that I was into electronics, everyone wanted me to repair and check their radios. I then made a big mistake of modifying a normal 2-chan radio to 3-chan so that at least you could fly with rudder, elevator and throttle control on a dihedral wing craft. I added another pot with a lever and some extra circuitry to the transmitter as well as a decoder expansion with some 4013 and 4017
  9. If you are referring to the bode plots of the PSU loop stability, that was injected with a current transformer inserted into the feedback path. I could have injected in the error amp's own feedback path as preferred by some people. In this case I don't think it will make much difference.
  10. Hi Omni, I used to have a Quad tube amp that sounded great. Only 15W rms but outperformed the 50 W transistor amps. Today I am sorry that I traded it for an old RF Motorola signal generator back in high school.
  11. Future Electronics have 10,000 + in stock at $0.18 ea. FC shows it is still in full production http://www.fairchildsemi.com/pf/MJ/MJE180.html
  12. I think I found a possible replacement. Look at the MJE180 from Fairchild. http://www.fairchildsemi.com/ds/MJ/MJE182.pdf Good, Ft (50 MHz), gain (50 min) and Ic (3A) Initial simulations look very good. Very stable with low ESR values and very fast loop load-step response (200nS). I will analyze this further, but it looks good. Mouser have them for $0.60 ea.
  13. That was the intention in the original design with a much better transistor. The whole picture changed. The situation will be the most problematic during load switching of high currents. It will be the worst at high output voltage settings and will get worse with lower ESR values. As-is, a 10uF cap with any ESR lower than 5 ohm will cause oscillation when switching.
  14. Sorry, Walid I will respond more in detail when I have more time.
  15. Plots for BD139 with cap and 2N2219 transistor
  16. Let's analyze the TIP31A from a S-parameter view as measured on a network analyzer using the actual components . We will look at the response of it operating as a simple emitter follower alone with only resistive terminations. This is normally considered to be a very stable configuration. Surprise! :o Good paper here for IEEE members http://ieeexplore.ieee.org/xpl/freeabs_all.jsp?tp=&arnumber=1082247&isnumber=23381 Looking at the first plot of a sweep between 0-10 MHz we see that this transistor is unstable with a K factor < 1 for operation above 50 kHz all the way up to 10 MHz. This transistor will oscillate without compensation. We normally aim for a K factor of at least >= 1.2 We can see a very good comparison between the measured S-param data and the model used by the simulator. We can thus assume the model is quite accurate. Next we add a common 10uF capacitor with an ESR of 7 ohm on the output. Now we see that this setup is unstable for anything over 70 kHz Next we take a look at the BD139 in the same configuration. This will be stable under 1.9 MHz (big difference from 50 kHz). Adding the same output cap we can see stability is not much affected with ample stability at 1 MHz (K = 1.9). Looking at the plot of the original 2N2219 we see a similar trend. The original designer may have had good reason for selecting the driver he did. Maybe he was aware of the problems like I am. The importance to select a fast driver is just as important in PSU design as in audio amplifiers. I can't make it any clearer than this.
  17. Let's look at a typical hookup using two 2N4444's in series. http://pdf1.alldatasheet.co.kr/datasheet-pdf/view/87010/MOTOROLA/2N4444.html Each has a blocking capability of 700V. We want to use it in a 1000V application. We connect it as shown in the first diagram, but find that the load turns on when we connect the power, even with no trigger applied. The problem is the leakage current that may vary from 10uA at a cold junction temperature to 2mA for a hot junction. In the diagram I used a leakage for 10uA for the one and about 20uA for the other. We can see that D1 with the lowest leakage will have more than its rated 700V over it. This will cause voltage breakdown of D1, followed by breakdown of D2. We need to create a additional leakage path greater that the worst leakage spec in order to balance the voltages. If we take 3mA to be very safe, the voltages balance out but we have to live with the high leakage current through the load, which may be a problem for the application. Also note that we will be generating a fair amount of heat in the balancing resistors as shown on the wattmeter.
  18. Look at this AP-note http://www.eetchina.com/ARTICLES/2000DEC/2000DEC12_AMD_AN5.PDF This gives you the basic idea, but there are other problem issues not mentioned here.
  19. Yes, with so many pirated and cloned parts around today you never know what you will get!
  20. Rather think of negative resistance as a negative differential resistance that behaves opposite to ohms law in that region. Some high frequency diodes also have this property, with the tunnel diode the most well known. Uni-junction transistors also operate on this principle. Another configuration you may want to lookup is the "lambda diode" that can be constructed with JFET's or a JFET-bipolar combination.
  21. The BD139 well heat-sinked will be a better choice with none of these problems. One can even look at a better output transistor with better gain to remove power issues in the BD139. It would be nice if we can get something with 2A capability and response of the 139
  22. Increasing phase lead will improve phase margin but at the same time will increase overshoot and ringing badly on a load step. Increasing the value to 330pF shows a 2V overshoot with lots of ringing. At 560pF it is at the point of oscillation with 2.5V overshoot. The oscillation problem just happen a bit higher in frequency. Increase it too much, say 1nF and we get constant oscillation at about 230 kHz, 1Vp-p at a constant full load.
  23. I agree simulators and models are not perfect but it shows a trend of what to expect. The models will tend to be much more accurate at fairly low frequencies than at the high end, where nonlinear models would be preferred. The problem might not be obvious as most members are using the cheap grade of output electrolytics with high ESR (1-10+ ohm for C7 ) values. The output may not oscillate but will have lousy load step performance (1A load step will drop 1-10V before the loop catches up). About the bodes, yes I am also used to the conventional form but the software does not display gain and phase at the same time, which is silly. The pole created by the TIP31A is the very same problem that trouble audio power amplifier design. If you have a slow driver stage you are looking at troubles. The slower output power stage does not make much additional phase difference. Doing the same simulation with any other slow (compared to the 2N2219 or BD139) transistor show similar problems. I have tried a few TIP's and also some of the higher current BD's like 241,243 and they present the same problem. We can blame the model, but all of them?
  24. This is what worries me. The plots compares the phase margins for: 1) Error amp alone with feedback network, no driver no filter cap C7 2) Error amp, with FB network and driver, no filter cap 3) Error amp, with FB, driver and filter cap 4) Error amp, with FB, driver, 2N3055 and filter cap Phase margin with the TIP31A driver is too small. Any variation in output cap ESR will bump this into oscillation at around 20-30 kHz. Using a quality low ESR capacitor for C7 to improve ESR voltage drop during current load step will be a disaster as the bode plot shows.
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